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LTC3542-1 500mA, 2.25MHz 2.8V Output Synchronous Step-Down DC/DC Converter FEATURES
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DESCRIPTION
The LTC(R)3542-1 is a high efficiency, fixed output voltage of 2.8V, monolithic synchronous buck converter using a constant frequency, current mode architecture. Supply current during operation is only 26A, dropping to <1A in shutdown. The 2.5V to 5.5V input voltage range makes the LTC3542-1 ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation, extending battery life in portable systems. Internal power switches are optimized to provide high efficiency and eliminate the need for an external Schottky diode. Switching frequency is internally set at 2.25MHz, allowing the use of small surface mount inductors and capacitors, and it can synchronize to an external clock signal with a frequency range of 1MHz to 3MHz through the MODE/SYNC pin. The LTC3542-1 is specifically designed to work well with ceramic output capacitors, achieving very low output voltage ripple and a small PCB footprint. The LTC3542-1 can be configured for the power saving Burst Mode(R) Operation. For reduced noise and RF interference, the MODE/SYNC pin can be configured for pulse skipping operation.
L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents including 5481178, 6580258, 6304066, 6127815, 6498466, 6611131, 5994885.
High Efficiency: Up to 96% Fixed Output Voltage: 2.8V High Peak Switch Current: 1000mA Low Output Ripple (<20mVP-P Typical) Burst Mode Operation Very Low Quiescent Current: Only 26A 2.5V to 5.5V Input Voltage Range 2.25MHz Constant Frequency Operation 1MHz to 3MHz External Frequency Synchronization Low Dropout Operation: 100% Duty Cycle No Schottky Diode Required Internal Soft-Start Limits Inrush Current Shutdown Mode Draws <1A Supply Current 2% Output Voltage Accuracy Current Mode Operation for Excellent Line and Load Transient Response Overtemperature Protected Available in 6-Lead 2mm x 2mm DFN
APPLICATIONS
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Cellular Telephones Wireless and DSL Modems Digital Cameras MP3 Players PDAs and Other Handheld Devices
TYPICAL APPLICATION
VIN 3V TO 5.5V 10F 2.2H VIN SW LTC3542-1 RUN VOUT VOUT 2.8V 500mA COUT 10F CER
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Efficiency and Power Loss vs Output Current
100 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 0.1 1 VIN = 3.6V VOUT = 2.8V 10 100 LOAD CURRENT (mA) 1 10 100 POWER LOSS (mW) 1000
MODE/SYNC GND
0.1 1000
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1
LTC3542-1 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW VOUT 1 VIN 2 GND 3 7 6 RUN 5 MODE/SYNC 4 SW
Input Supply Voltage (VIN) ........................... -0.3V to 6V VOUT, RUN Voltages......................................-0.3V to VIN MODE Voltage ................................-0.3V to (VIN + 0.3V) SW Voltage ....................................-0.3V to (VIN + 0.3V) Operating Temperature Range (Note 2).... -40C to 85C Junction Temperature (Note 7) ............................. 125C Storage Temperature Range................... -65C to 150C Reflow Peak Body Temperature ............................ 260C
DC PACKAGE 6-LEAD (2mm 2mm) PLASTIC DFN TJMAX = 125C, JA = 89C/W, JC = 18C/W (SOLDERED TO A 4-LAYER BOARD, NOTE 3) EXPOSED PAD (PIN 7) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3542EDC-1#PBF TAPE AND REEL LTC3542EDC-1#TRPBF PART MARKING LDWC PACKAGE DESCRIPTION 6-Lead (2mm x 2mm) Plastic DFN TEMPERATURE RANGE -40C to 85C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V unless otherwise noted.
SYMBOL VIN RVOUT VOUT VLINE_REG VLOAD_REG IS PARAMETER Operating Voltage Range Input Resistance of VOUT Pin Output Feedback Voltage (Note 4) Reference Voltage Line Regulation (Note 4) VIN = 2.5V to 5.5V Output Voltage Load Regulation (Note 4) Input DC Supply Current (Note 5) Active Mode Sleep Mode Shutdown Oscillator Frequency Synchronous Frequency Peak Switch Current P-Channel On Resistance (Note 6) N-Channel On Resistance (Note 6) Switch Leakage Current ILOAD = 100mA to 500mA VOUT = 2.5V VOUT = 2.9V, MODE = 0V RUN = 0V VOUT = 2.8V VOUT = 2.8V VIN = 3V, VOUT = 2.5V, Duty Cycle < 35% ISW = 100mA ISW = -100mA VIN = 5V, VRUN = 0V, VSW = 0V or 5V
l l
ELECTRICAL CHARACTERISTICS
CONDITIONS
l
MIN 2.5 504 2.744
TYP 840 2.8 0.04 0.02 300 26 0.1
MAX 5.5 1176 2.856 0.2 0.2 500 35 1 2.7 3
UNITS V k V %/V % A A A MHz MHz mA A
fOSC fSYNC ILIM RDS(ON) ISW(LKG)
1.8 1 650
2.25 1000 0.5 0.35 0.01
0.65 0.55 1
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LTC3542-1 ELECTRICAL CHARACTERISTICS
SYMBOL VUVLO VRUN IRUN VMODE/SYNC IMODE/SYNC PARAMETER Undervoltage Lockout Threshold RUN Threshold RUN Leakage Current MODE/SYNC Threshold MODE/SYNC Leakage Current
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V unless otherwise noted.
CONDITIONS VIN Rising VIN Falling
l l l l
MIN 1.8 0.3
TYP 2.0 1.9 0.01
MAX 2.3 1.5 1 1.2 1
UNITS V V V A V A
0.3 0.01
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. No pin should exceed 6V. Note 2: The LTC3542-1 is guaranteed to meet performance specifications from 0C to 85C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Failure to solder the Exposed Pad of the package to the PC board will result in a thermal resistance much higher than 89C/W. Note 4: The converter is tested in a proprietary test mode that connects
the output of the error amplifier to the SW pin, which is connected to an external servo loop. Note 5: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. Note 6: The DFN switch on resistance is guaranteed by correlation to wafer level measurements. Note 7: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD) * (JA).
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LTC3542-1 TYPICAL PERFORMANCE CHARACTERISTICS
Burst Mode Operation TA = 25C unless otherwise specified. Start-Up from Shutdown
Pulse-Skip Mode Operation
SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 100mA/DIV VIN = 3.6V 2s/DIV ILOAD = 25mA FIGURE 3a CIRCUIT
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SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 100mA/DIV
RUN 2V/DIV VOUT 1V/DIV
IL 200mA/DIV VIN = 4.2V ILOAD = 0A 400ns/DIV
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400s/DIV VIN = 3.6V ILOAD = 0A FIGURE 3a CIRCUIT
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Start-Up from Shutdown
VOUT 100mV/DIV AC COUPLED IL 500mA/DIV
Load Step
VOUT 100mV/DIV AC COUPLED IL 500mA/DIV
Load Step
RUN 2V/DIV VOUT 1V/DIV IL 200mA/DIV
ILOAD 500mA/DIV VIN = 3.6V 400s/DIV ILOAD = 500mA FIGURE 3a CIRCUIT
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ILOAD 500mA/DIV VIN = 3.6V 20s/DIV ILOAD = 30mA TO 500mA FIGURE 3a CIRCUIT
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VIN = 3.6V 20s/DIV ILOAD = 0mA TO 500mA FIGURE 3a CIRCUIT
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Reference Voltage vs Temperature
2.85 VIN = 3.6V PULSE-SKIP MODE NO LOAD FREQUENCY (MHz) 2.7 2.6 2.5
Oscillator Frequency vs Temperature
2.7 2.6 2.5 FREQUENCY (MHz) 2.4 2.3 2.2 2.1 2.0 1.9 1.8 90 110
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Oscillator Frequency vs Supply Voltage
2.83
2.4 2.3 2.2 2.1 2.0 1.9
VOUT (V)
2.81
2.79
2.77
2.75 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
1.8 -50 -30 -10 10 30 50 70 TEMPERATURE (C)
2
3
5 4 SUPPLY VOLTAGE (V)
6
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LTC1520 G01
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LTC3542-1 TYPICAL PERFORMANCE CHARACTERISTICS
Output Voltage vs Supply Voltage
1.0 0.8 0.6 VOUT ERROR (%) VOUT ERROR (%) 0.4 0.2 0 1.0 RDS(ON) () 0.5 0 PULSE SKIP -0.5 -1.0 -1.5 -2.0 2 3 4 5 INPUT VOLTAGE (V) 6
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TA = 25C unless otherwise specified. RDS(ON) vs Input Voltage
0.9 0.8 0.7 0.6
Output Voltage vs Load Current
2.0 1.5 VIN = 3.6V VOUT = 2.8V
IOUT = 100mA FIGURE 3a CIRCUIT
Burst Mode OPERATION
MAIN SWITCH 0.5 0.4 0.3 0.2 0.1 0 SYNCHRONOUS SWITCH
-0.2 -0.4
-0.6 -0.8 -1.0
1
10 100 LOAD CURRENT (mA)
1000
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1
2
3
4 VIN (V)
5
6
7
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RDS(ON) vs Temperature
0.9 0.8 LEAKAGE CURRENT (pA) 0.7 0.6 RDS(ON) () 0.5 0.4 0.3 0.2 0.1 0 -50 -25 0 25 50 75 TEMPERATURE (C) VIN = 3.6V VIN = 4.2V 100 125 SYNCHRONOUS SWITCH MAIN SWITCH 1000 900 800 700 600 500 400 300 200 100 0
Switch Leakage vs Input Voltage
300 250 SWITCH LEAKAGE (nA) 200 150 100 50
Switch Leakage vs Temperature
MAIN SWITCH
SYNCHRONOUS SWITCH
MAIN SWITCH SYNCHRONOUS SWITCH 50 25 75 0 TEMPERATURE (C) 100 125
0
1
2
3 VIN (V)
4
5
6
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0 -50 -25
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Efficiency vs Input Voltage
100 90 80 EFFICIENCY (%) EFFICIENCY (%) 70 60 50 40 30 3 IOUT = 0.1mA IOUT = 1mA IOUT = 10mA IOUT = 100mA IOUT = 500mA 3.5 5 4.5 4 INPUT VOLTAGE (V) 5.5 6 100 90 80 70 60 50 40 30 20 10
Efficiency vs Load Current
VIN = 3.6V VIN = 4.2V EFFICIENCY (%) 100
Efficiency vs Load Current
VIN = 3.6V VOUT = 2.8V
90
80
Burst Mode OPERATION
PULSE SKIP
70
60
FIGURE 3a CIRCUIT 0 0.1 1 10 100 LOAD CURRENT (mA)
1000
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50 0.1
1
10 100 LOAD CURRENT (mA)
1000
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LTC3542-1 PIN FUNCTIONS
VOUT (Pin 1): Output Pin. Receives the 2.8V output voltage to internal feedback resistors. VIN (Pin 2): Power Supply Pin. Must be closely decoupled to GND. GND (Pin 3): Ground Pin. SW (Pin 4): Switch Node Connection to Inductor. This pin connects to the drains of the internal main and synchronous power MOSFET switches. MODE/SYNC (Pin 5): Mode Selection and Oscillator Synchronization Pin. This pin controls the operation of the device. When tied to GND or VIN, Burst Mode operation or pulse skipping mode is selected, respectively. The oscillation frequency can be synchronized to an external oscillator applied to this pin and pulse skipping mode is automatically selected. Do not float this pin. RUN (Pin 6): Converter Enable Pin. Forcing this pin above 1.5V enables this part, while forcing it below 0.3V causes the device to shut down. In shutdown, all functions are disabled drawing <1A supply current. This pin must be driven; do not float. GND (Pin 7): Exposed Pad. The Exposed Pad is ground. It must be soldered to PCB ground to provide both electrical contact and optimum thermal performance.
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LTC3542-1 BLOCK DIAGRAM
SLOPE COMPENSATION OSC ICOMP VOUT R1 R2 0.6V
+ -
VIN
EA
VB
+
MODE/SYNC VIN
RUN
0.6V REF
SHUTDOWN
+
-
BURST MODE MODE DETECT CLKIN LOGIC ANTISHOOT THROUGH SW
+ +
IRCMP
-
-
GND
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LTC3542-1 OPERATION
The LTC3542-1 uses a constant frequency, current mode, step-down architecture. The operating frequency is set at 2.25MHz and can be synchronized to an external oscillator. To suit a variety of applications, the selectable MODE/SYNC pin allows the user to trade off noise for efficiency. The output voltage is set by an internal resistor divider. An error amplifier compares the divided output voltage with a reference voltage of 0.6V and adjusts the peak inductor current accordingly. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle when the divided output voltage is below the reference voltage. The current flows into the inductor and the load increases until the current limit is reached. The switch turns off and energy stored in the inductor flows through the bottom switch (N-channel MOSFET) into the load until the next clock cycle. The peak inductor current is controlled by the internally compensated output of the error amplifier. When the load current increases, the divided output voltage decreases slightly below the reference. This decrease causes the error amplifier to increase its output voltage until the average inductor current matches the new load current. The main control loop is shut down by pulling the RUN pin to ground. Low Load Current Operation By selecting MODE/SYNC pin, two modes are available to control the operation of the LTC3542-1 at low load currents. Both modes automatically switch from continuous operation to the selected mode when the load current is low. To optimize efficiency, the Burst Mode operation can be selected. When the converter is in Burst Mode operation, the peak current of the inductor is set to approximately 125mA regardless of the output load. Each burst event can last from a few cycles at light loads to almost continuously cycling with short sleep intervals at moderate loads. In between these burst events, the power MOSFETs and any unneeded circuitry are turned off, reducing the quiescent current to 26A. In this sleep state, the load current is being supplied solely from the output capacitor. As the output voltage drops, the EA amplifier's output rises above the sleep threshold and turns the top MOSFET on. This process repeats at a rate that is dependent on the load demand. By running cycles periodically, the switching losses which are dominated by the gate charge losses of the power MOSFETs are minimized. For lower ripple noise at low load currents, the pulse skip mode can be used. In this mode, the regulator continues to switch at a constant frequency down to very low load currents, where it will begin skipping pulses. Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases to 100%, which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. An important design consideration is that the RDS(ON) of the P-channel switch increases with decreasing input supply voltage (See Typical Performance Characteristics). Therefore, the user should calculate the power dissipation when the LTC3542-1 is used at 100% duty cycle with low input voltage (See Thermal Considerations in the Applications Information Section). Low Supply Operation To prevent unstable operation, the LTC3542-1 incorporates an undervoltage lockout circuit which shuts down the part when the input voltage drops below about 2V. Internal Soft-Start At start-up when the RUN pin is brought high, the internal reference is linearly ramped from 0V to 0.6V in about 1ms. The regulated feedback voltage follows this ramp resulting in the output voltage ramping from 0% to 100% in 1ms. The current in the inductor during soft-start is defined by the combination of the current needed to charge the output capacitance and the current provided to the load as the output voltage ramps up. The start-up waveform, shown in the Typical Performance Characteristics, shows the output voltage start-up from 0V to 2.8V with a 500mA load and VIN = 3.6V (refer to Figure 3a).
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LTC3542-1 APPLICATIONS INFORMATION
A general LTC3542-1 application circuit is shown in Figure1. External component selection is driven by the load requirement and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected.
L VIN CIN VIN RUN SW VOUT VOUT COUT
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the burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance values cause the burst frequency to increase. Inductor Core Selection Different core materials and shapes change the size/current and price/current relationships of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on what the LTC3542-1 requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3542-1 applications. Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT/VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT ( VIN - VOUT ) VIN
LTC3542-1 MODE/SYNC GND
Figure 1. LTC3542-1 General Schematic
Inductor Selection The inductor value has a direct effect on ripple current IL, which decreases with higher inductance and increases with higher VIN or VOUT , as shown in following equation: IL = VOUT VOUT 1- O * L VIN
where fO is the switching frequency. A reasonable starting point for setting ripple current is IL = 0.4 * IOUT(MAX), where IOUT(MAX) is 500mA. The largest ripple current IL occurs at the maximum input voltage. To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation: L= VOUT VOUT 1- O * IL VIN(MAX )
The DC current rating of the inductor should be at least equal to the maximum load current plus half the ripple current to prevent core saturation. Thus, a 600mA rated inductor should be enough for most applications (500mA + 100mA). For better efficiency, chose a low DC-resistance inductor. The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the inductor's peak current falls below a level set by
where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - IL/2. This formula has a maximum at VIN = 2VOUT , where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours life time. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the
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LTC3542-1 APPLICATIONS INFORMATION
Table 1. Representative Surface Mount Inductors
VALUE (H) 2.2 2.2 2.2 3.3 4.7 2.2 4.7 2.2 2.2 MAX DC CURRENT (A) 0.780 1.2 0.95 0.85 0.75 0.79 0.75 0.85 1.0 DCR () 0.098 0.075 0.116 0.15 0.15 0.15 0.18 0.12 SIZE (mm3) 3.2 x 3.2 x 1.2 3.8 x 3.8 x 1.8 4.4 x 5.8 x 1.2 2.8 x 3 x 1 4.9 x 4.9 x 1 4.5 x 3.2 x 2.6 4.8 x 3.4 x 3.4 2.8 x 2.6 x 1
MANUFACTURER Sumida
PART NUMBER CDRH2D11-2RM CDRH3D16 CMD4D11 CDH2D09B CLS4D09
Murata TDK
LQH32CN LQH43CN IVLC453232 VLF3010AT2R2M1R0
0.097 2.5 x 3.2 x 1.55
design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling, when not using an all ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required ESR to minimize voltage ripple and load step transients. Typically, once the ESR requirement is satisfied, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement, except for an all ceramic solution. The output ripple (VOUT) is determined by: 1 VOUT IL ESR + 8 * O * COUT where fO is the switching frequency, COUT is the output capacitance and IL is the inductor ripple current. For a fixed output voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP Kemet T510 and ,
T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current rating, high voltage rating and low ESR are tempting for switching regulator use. However, the ESR is so low that it can cause loop stability problems. Since the LTC3542-1's control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used to achieve very low output ripple and small circuit size. X5R or X7R ceramic capacitors are recommended because these dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Great care must be taken when using only ceramic input and output capacitors. When a ceramic capacitor is used at the input and the power is being supplied through long wires, such as from a wall adapter, a load step at the output can induce ringing at the VIN pin. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, the ringing at the input can be large enough to damage the part. For more information, see Application Note 88. The recommended capacitance value to use is 10F for both input and output capacitors.
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LTC3542-1 APPLICATIONS INFORMATION
Mode Selection and Frequency Synchronization The MODE/SYNC pin is a multipurpose pin that provides mode selection and frequency synchronization. Connecting this pin to GND enables Burst Mode operation, which provides the best low current efficiency at the cost of a higher output voltage ripple. Connecting this pin to VIN selects pulse skip mode operation, which provides the lowest output ripple at the cost of low current efficiency. The LTC3542-1 can also be synchronized to an external clock signal with range from 1MHz to 3MHz by the MODE/SYNC pin. During synchronization, the mode is set to pulse skip and the top switch turn-on is synchronized to the falling edge of the external clock. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, three main sources usually account for most of
1000
the losses in LTC3542-1 circuits: 1) VIN quiescent current, 2) I2R loss and 3) switching loss. VIN quiescent current loss dominates the power loss at very low load currents, whereas the other two dominate at medium to high load currents. In a typical efficiency plot, the efficiency curve at very low load currents can be misleading since the actual power loss is of no consequence as illustrated in Figure 2. 1) The VIN quiescent current is the DC supply current given in the Electrical Characteristics which excludes MOSFET charging current. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 2) I2R losses are calculated from the DC resistances of the internal switches, RSW, and external inductor, RL. In continuous mode, the average output current flows through inductor L, but is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (D) as follows: RSW = (RDS(ON)TOP)(D) + (RDS(ON)BOT)(1 - D) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2(RSW + RL)
VIN = 3.6V FIGURE 3a CIRCUIT
POWER LOSS (mW)
10
0.1 0.1
1
10 100 LOAD CURRENT (mA)
1000
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Figure 2. Power Loss vs Load Current
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LTC3542-1 APPLICATIONS INFORMATION
3) The switching current is MOSFET gate charging current, that results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. Other "hidden" losses such as copper trace and internal battery resistances can account for additional efficiency degradations in portable systems. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses include diode conduction losses during dead-time and inductor core losses generally account for less than 2% total additional loss. Thermal Considerations In most applications the LTC3542-1 does not dissipate much heat due to its high efficiency. But in applications where the LTC3542-1 is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 160C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3542-1 from exceeding the maximum junction temperature, the user need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = (PD)(JA) where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient. The junction temperature, TJ, is given by: TJ = TA + TR where TA is the ambient temperature. As an example, consider the LTC3542-1 at an input voltage of 3.6V, a load current of 500mA and an ambient temperature of 70C. From the typical performance graph of switch resistance, the RDS(ON) of the P-channel switch at 70C is approximately 0.6. Therefore, power dissipated by the part is: PD = ILOAD2 * RDS(ON) = 150mW For the DFN package, the JA is 89C/W. Thus, the junction temperature of the regulator is: TJ = 70C + 0.150 * 89 = 83.4C which is below the maximum junction temperature of 125C. Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)). Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD * ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT, generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. In some applications, a more severe transient can be caused by switching loads with large (>1F) bypass capacitors. The discharged bypass capacitors are effectively put in
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LTC3542-1 APPLICATIONS INFORMATION
parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limit, short circuit protection and soft-start. Design Example As a design example, assume the LTC3542-1 is used in a single lithium-ion battery-powered cellular phone application. The VIN will be operating from a maximum of 4.2V down to 2.8V. The load current requirement is a maximum of 0.5A, but most of the time it will be in standby mode, requiring only 2mA. Efficiency at both low and high load currents is important. Output voltage is 2.8V. With this information we can calculate L using: V 1 L= * VOUT * 1- OUT f * IL VIN Substituting VOUT = 2.8V, VIN = 4.2V, IL = 200mA and f = 2.25MHz gives: 2.8 V 2.8 V L= * 1- = 2.07H 2.25MHz * 200mA 4.2V Choosing a vendor's closest inductor value of 2.2H results in a maximum ripple current of: IL = 2.8 V 2.8 V * 1 - 6 = 188.6mA 2.25MHz * 2.2H 4.2V CIN will require an RMS current rating of at least 0.25A ILOAD(MAX)/2 at temperature and COUT will require ESR of less than 0.2. In most cases, ceramic capacitors will satisfy these requirements. Select COUT = 10F and . CIN = 10F Figure 3 shows the complete circuit along with its efficiency curve, load step response and recommended layout. PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3542-1. These items are also illustrated graphically in Figure 3b. Check the following in your layout: 1. The power traces, consisting of the GND trace, the SW trace and the VIN trace should be kept short, direct and wide. 2. Does the VOUT pin connect directly to the (+) plate of COUT? 3. Does the (+) plate of CIN connect to VIN as closely as possible? This capacitor provides the AC current to the internal power MOSFETs. 4. Keep the (-) plates of CIN and COUT as close as possible.
Hot Swap is a trademark of Linear Technology Corporation.
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LTC3542-1 APPLICATIONS INFORMATION
VIN 3V TO 5.5V L* 2.2H VIN CIN** 10F SW LTC3542-1 RUN VOUT COUT** 10F
35421 F03a
VOUT 2.8V 500mA
MODE/SYNC GND
*SUMIDA CDRH2D18HP-2R2NC **TDK C2012X5R0J106M
Figure 3a. Typical Application
GND
VIN
VOUT 1 VIN 2 CIN GND 3 GND
6 RUN 5 MODE/ SYNC 4 SW
VIA TO VOUT
L GND COUT
VOUT
35421 F03b
Figure 3b. Layout Diagram
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 FIGURE 3a CIRCUIT 0 0.1 1 10 100 LOAD CURRENT (mA) 1000 VIN = 3.6V 20s/DIV ILOAD = 30mA TO 500mA FIGURE 3a CIRCUIT
35421 G05
VIN = 3.6V
VOUT 100mV/DIV AC COUPLED IL 500mA/DIV
ILOAD 500mA/DIV
Figure 3d. Load Step
35421 F03c
Figure 3c. Efficiency Curve
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LTC3542-1 PACKAGE DESCRIPTION
DC Package 6-Lead Plastic DFN (2mm x 2mm)
(Reference LTC DWG # 05-08-1703)
0.675 0.05 2.50 0.05 1.15 0.05 0.61 0.05 (2 SIDES) PACKAGE OUTLINE
0.25 0.50 BSC 1.42 0.05 (2 SIDES)
0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS R = 0.115 TYP 0.56 0.05 (2 SIDES) 2.00 0.10 (4 SIDES) PIN 1 CHAMFER OF EXPOSED PAD 3 0.200 REF 0.75 0.05 1 0.25 0.50 BSC 1.37 0.05 (2 SIDES) 0.00 - 0.05 BOTTOM VIEW--EXPOSED PAD
(DC6) DFN 1103
0.38 4 6
0.05
PIN 1 BAR TOP MARK (SEE NOTE 6)
0.05
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WCCD-2) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LTC3542-1 TYPICAL APPLICATION
Using Low Profile Components, <1mm Height
VIN 3V TO 5.5V 2 CIN** 10F CER 6 5 VIN RUN SW VOUT 4 1 COUT** 10F CER
35421 TA02a
2.2H*
LTC3542-1 MODE/SYNC GND 3
VOUT 2.8V 500mA
*TDK VLF3010AT-2R2MIR0 **TDK C2012X5R0J106M
Efficiency vs Load Current
100
90 EFFICIENCY (%)
80
70
60
50 0.1
VIN = 3.6V VOUT = 2.8V Burst Mode OPERATION 1 10 100 LOAD CURRENT (mA) 1000
35421 TA02b
RELATED PARTS
PART NUMBER LTC3405/LTC3405B LTC3406/LTC3406B LTC3407/LTC3407-2 LTC3409 LTC3410/LTC3410B LTC3411 LTC3548 LTC3561 DESCRIPTION 300mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC Converter 600mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC Converter COMMENTS 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 20A, ISD < 1A, ThinSOT Package 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20A, ISD < 1A, ThinSOT Package
Dual 600mA/800mA IOUT, 1.5MHz/2.25MHz, Synchronous 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, Step-Down DC/DC Converter ISD < 1A, MS10E, DFN Packages 600mA IOUT, 1.7MHz/2.6MHZ, Synchronous Step-Down DC/DC Converter 300mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter 1.25A IOUT, 4MHz, Synchronous Step-Down DC/DC Converter Dual 400mA/800mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter 1A IOUT, 4MHz Synchronous Step-Down DC/DC Converter 96% Efficiency, VIN: 1.6V to 5.5V, VOUT(MIN) = 0.6V, IQ = 65A, ISD < 1A, DFN Package 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26A, ISD < 1A, SC70 Package 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD < 1A, MS10, DFN Packages 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD < 1A, MS10, DFN Packages 95% Efficiency, VIN: 2.6V to 5.5V, VOUT(MIN) = 0.8V, IQ = 240A, ISD < 1A, 3mm x 3mm DFN Package
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
LT 0708 * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2008

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